FT021F HX1304F HX13C HC2018 HC2018A

产品特点宽输入电压范围：１０－３０Ｖ高达９３％的效率

可编程开关频率高达高达５００ｋＨｚ无环路补偿可编程电流电缆补偿从０到０．６热关断

电流模式能提供快速瞬态响应和简化环路稳定性 ＦＴ０２１Ｆ　ＰＣＢＡ版上外部极少元器件

其他功能还包括电压补偿，可调制电流和热关断保护

该ＦＴ０２１Ｆ　ＤＣ－ＤＣ转换器　封装体积为ＳＯＰ－８Ｌ 产品应用车载充电器／适配器

预调节的线性稳压器分布式电源系统电池充电器

TOP VIEW

S/W 典型应用电路10

10015031.6

33UHFT021F ＦＴ０２１Ｆ100200

* The output voltage is set by R2 and R3: VOUT = 1.21V • [1 + (R2/R3)].

Pin Assignment and Description

PIN

NAME

DESCRIPTION Feedback Frequency Setting Current Limit Input Supply Voltage

Switch Node Ground

1 FB 2 RT 3 ILIM 4 VIN 5, 6 7, 8

SW GND

Absolute Maximum Ratings (Note 1)

 Input Supply Voltage ....................................................................................................-0.3V ～ 35V  FB, ILIM, RT Voltages.................................................................................................... -0.3V ～ 6V  SW Voltage ........................................................................................................-0.3V ～ (VIN + 1V)  Operating Temperature Range (Note 2)………...………………………………………-40℃ ～ +85℃  Storage Temperature Range.................................................................................. -65℃ ～ +150℃  Junction Temperature Range………………………………………………...……………..……...+150℃  Lead Temperature (Soldering, 10 sec.).................................................................................. +265℃

注１：

超出所列的绝对最大额定值强调可能会造成永久性损坏设备。

于任何绝对最大额定值长时间状态可能会影响器件的可靠性和寿命。注２：

该ＦＴ０２１Ｆ是保证符合性能规格为０度－７０度，在－４０至＋８５度工作温度范围内放心通过设计，表征和相关的统计过程控制。

FT021FElectrical Characteristics

Operating Conditions: TA=25℃, VIN=12V, R2=470k, R3=150k, unless otherwise specified. SYMBOL VIN

PARAMETER

Operating Voltage Range

CONDITIONS

MIN 10

TYP

MAX 30

UNITSV

IQ Quiescent Current =ILOAD0A 10 15 20 mA ISHDN Shutdown Current VUVLO

Input UVLO Threshold

110 150 μA

4.25 4.5 V ΔVUVLO UVLO Hysteresis VFB Regulated Voltage IFB fOSC DC ILIM-TH RPFET RNFET TSD ΔTSD

Feedback Pin Input Current

50 100 mV

1.188 1.21 1.236 V 0.05 μA

100 500 kHz Oscillator Frequency Range

=RT100k 180 220 260 kHz Max Duty Cycle Current Limit Sense Pin

Source Current

RDS(ON) of P-Channel FET RDS(ON) of N-Channel FET Thermal Shutdown Thermal Shutdown Hysteresis

100

%

7 8.5 10 μA

65 mΩ

30 mΩ Temperature Rising

150 ℃

30 ℃

FT021FTypical Performance Characteristics

Operating Conditions: TA=25℃, CIN=47μF, COUT=100μF, L=10μH, unless otherwise noted.

Vin=24V

Vin=12V

FT021FPin Functions

FB (Pin 1): Feedback Pin. Receive the feedback voltage from an external resistive divider across the output. In the adjustable version, the output voltage is fixed. The Output voltage is set by R2 and R3: VOUT = 1.21V • [1 + (R2/R3)].

RT (Pin 2): The internal oscillator is set with a single resistor between this pin and the GND pin. ILIM (Pin 3): Monitors current through the low-side switch and triggers current limit operation if the inductor valley current exceeds a user defined value that is set by RLIM and the Sense current sourced out of this pin during operation.

VIN (Pin 4): Main Supply Pin. The FT021F operates fromfrom appearing at the input.

SW (Pin 5, 6): Switch Node Connection to Inductor. GND (Pin 7, 8): Ground Pin.

10V to 30V unregulated input. It must be

closely decoupled to GND, with a 47μF or greater ceramic capacitor to prevent large voltage spikes

Block Diagram

FT021FApplication Information

The FT021F operates by a

constant frequency, current mode architecture. The output voltage is set by

an external divider returned to the FB pin. An error amplifier compares the divided output voltage with a reference voltage of 1.21V and adjusts the peak inductor current accordingly.

During normal operation, the internal P-channel MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, resets the RS latch. While the P-channel MOSFET is off, the N-channel MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator or the beginning of the next clock cycle. Thermal Protection

The total power dissipation in FT021F is limited

by a thermal protection circuit. When the device

temperature rises to approximately 150℃, this circuit turns off the output, allowing the IC to cool. The thermal protection circuit can protect the device from being damaged by overheating in the event of fault conditions. Continuously running the FT021FCurrent Limit

Current limit detection occurs during the off-time by monitoring the current through the low-side switch using an external resistor, RLIM. The current limit value is defined by RLIM. If during the off-time the current in the low-side switch exceeds the user defined current limit value, the next on-time cycle is immediately terminated. Current sensing is achieved by comparing the voltage across the low side FET with the voltage across the current limit set resistor RLIM. For example, the current limit value is 2.5A by the RLIM =62k. The current limit value rises when the set resistor RLIM rises. The maximum output current is set by RLIM: RLIM (kΩ) = 24• IMAX (A). Oscillator Frequency

The FT021F oscillator frequency is set by a single

into thermal shutdown degrades device reliability.

external resistor connected between the RT pin and

the GND pin. The resistor should be located very close to the device and connected directly to the pins of the IC (RT and GND). An internal amplifier holds the RT pin at a fixed voltage typically 0.6V. The oscillator frequency rises when the resistor RT falls. To determine the timing resistance for a given switching frequency, use the equation below:

RT(kΩ)= 22000 /fOSC(kHz)

Setting Output Voltage

The output voltage is set with a resistor divider from the output node to the FB pin. It is recommended to use divider resistors with 1% tolerance or better. To improve efficiency at very light loads consider using larger value resistors. If the values are too high the regulator is more susceptible to noise and voltage errors from the FB input current are noticeable. For most applications, a resistor in the 10kΩ to 1MΩ range is suggested for R3. R2 is then given by:

R2 = R3 • [(VOUT / VREF) – 1]

where VREF is 1.21V.

FT021FOutput Cable Resistance Compensation

To compensate for resistive voltage drop across the charger's output cable, the FT021F integrates a simple, user-programmable cable voltage drop compensation using the impedance at the FB pin. Choose the proper feedback resistance values for cable compensation refer to the curve in Figure 1. The delta VOUT voltage rises when the feedback resistance R3 value rises. The delta VOUT voltage rises when the feedback resistance R3 value rises, use the equation below:

ΔVOUT(V) = R3(kΩ) • IOUT(A)/2000

Figure 1. Delta Output Voltage vs. Load Current

Inductor Selection

For most applications, the value of the inductor will fall in the range of 4.7μH to 47μH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation. A reasonable starting point for setting ripple current is △IL=1A (40% of 2.5A).

The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 2.9A rated inductor should be enough for most applications (2.5A + 400mA). For better efficiency, choose a low DC-resistance inductor.

Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or perm alloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs. size requirements and any radiated field/EMI requirements than on what the FT021F requires to operate.

FT021FOutput and Input Capacitor Selection

In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:

This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question.

The selection of COUT is driven by the required effective series resistance (ESR).Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ΔVOUT is determined by:

Where f = operating frequency, COUT = output capacitance and ΔIL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ΔIL increases with input voltage.

Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Efficiency Considerations

The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1+ L2+ L3+ ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence.

1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge ΔQ moves from VIN to ground. The resulting ΔQ/Δt is the current out of VIN that is typically larger than the DC bias current.

FT021FIn continuous mode, IGATECHG = f (QT+QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages.

2. I2R losses are calculated from the resistances of the internal switches, RSW and external inductor RL. In continuous mode the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = RDS(ON)TOP x DC + RDS(ON)BOT x (1-DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% of the total loss.Board Layout Suggestions

When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the FT021F. Check the following in your layout.

1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept

short, direct and wide. 2. Put the input capacitor as close as possible to the device pins (VIN and GND).

3. SW node is with high frequency voltage swing and should be kept small area. Keep analog

components away from SW node to prevent stray capacitive noise pick-up. 4. Connect all analog grounds to a command node and then connect the command node to the power

ground behind the output capacitors.

FT021FPackaging Information

SOP-8L Package Outline Dimension

Symbol

Dimensions In Millimeters Min Max

Dimensions In Inches

Min Max

A 1.350 1.750 0.053 0.069 A1 0.100 A2 1.350 0.250 0.004 0.010 1.550 0.053 0.061 b 0.330 0.510 0.013 0.020 c 0.170 0.250 0.006 0.010 D 4.700 5.100 0.185 0.200 E 3.800 4.000 0.150 0.157 E1 5.800 6.200 0.228 0.244 0.050(BSC) e 1.270(BSC) θ 0°

L 0.400 1.270 0.016 0.050 8° 0° 8°